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MP1517
3A, 25V, 1.1MHz Step-Up Converter
The Future of Analog IC Technology
DESCRIPTION
The MP1517 is a 3A, fixed frequency step-up converter ideal for medium-to-high current step-up, flyback and SEPIC applications. The high 1.1MHz switching frequency allows for smaller external components producing a compact solution for size constrained cameras, PDAs and cell phones. The low 0.7V feedback voltage offers higher efficiency in white LED driver applications including cell phone camera flash. The MP1517 regulates the output voltage up to 25V with efficiencies as high as 95%. Soft-start, cycle-by-cycle current limiting, and input under voltage lockout prevent overstressing or damage to sensitive external circuitry at startup and output short-circuit conditions. Current-mode regulation and external compensation components allow the MP1517 control loop to be optimized over a wide variety of input voltage, output voltage, and load current conditions. The MP1517 is available in the thermally enhanced QFN16 (4mm x 4mm) package.
FEATURES
* * * * * * * * * * * * * * * 4A Peak Current Limit Low 700mV Feedback Threshold Internal 150m Power Switch Input Range of 2.6V to 25V Up to 95% Efficiency Zero Current Shutdown Mode Under Voltage Lockout Protection Open Load Protection Soft-Start Operation Thermal Shutdown Tiny 4mm x 4mm 16-Pin QFN Package Boost and SEPIC Regulators Handheld Computers Cell Phone Camera Flash, PDAs Digital Still and Video Cameras
APPLICATIONS
"MPS" and "The Future of Analog IC Technology" are Registered Trademarks of Monolithic Power Systems, Inc.
EVALUATION BOARD REFERENCE
Board Number EV0043 Dimensions 2.5"X x 2.0"Y x 0.4"Z
TYPICAL APPLICATION
VIN 5V
Efficiency vs Load Current
100
8 IN 4 15 9, 10
D1 MBR320
95
EFFICIENCY (%)
VOUT 12V
OFF ON
EN SS
SW
90 85 80 75 70 65 60 55 50 1
VOUT = 12V
C4 10nF C5 10nF
3 7
MP1517
BP OLS SGND 5, 13 FB
16 1
COMP PGND 11, 12
VOUT = 18V VIN = 5V 10 100 LOAD CURRENT (mA) 1000
MP1517-EC01
C3 10nF
C6 OPTIONAL
MP1517_TAC_S01
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER
PACKAGE REFERENCE
TOP VIEW
PIN 1 IDENTIFICATION FB 16 1 2 3 4 5 SGND 6 NC 7 OLS 8 IN
MP1517_PD01-QFN16
ABSOLUTE MAXIMUM RATINGS (1)
EXPOSED PAD CONNECT TO PIN 13
SS 15
NC 14
SGND 13 12 11 10 9
COMP NC BP EN
PGND PGND SW SW
Input Supply Voltage VIN .............. -0.3V to +27V SW Pin Voltage VSW .................... -0.3V to +27V Voltage at All Other Pins except OLS .................................-0.3V to +6V Storage Temperature ..............-65C to +150C Junction Temperature.............................+150C
Recommended Operating Conditions
(2)
IN Input Supply Voltage VIN ............. 2.6V to 25V Output Voltage................................. 3.3V to 25V Operating Temperature .............-40C to +85C
Thermal Resistance
(3)
Part Number* MP1517DR *
Package QFN16 (4mm x 4mm)
Temperature -40C to +85C
QFN16 (4mm x 4mm) ............. 46 ...... 10... C/W
Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1" square of 1 oz. Copper.
JA
JC
For Tape & Reel, add suffix -Z (eg. MP1517DR-Z) For Lead Free, add suffix -LF (eg. MP1517DR-LF-Z)
ELECTRICAL CHARACTERISTICS
VIN = 5.0V, TA = +25C, unless otherwise noted.
Parameter IN Shutdown Supply Current IN Operating Supply Current BP Voltage IN Undervoltage Lockout Threshold IN Undervoltage Lockout Hysteresis EN Input Low Voltage EN Input High Voltage EN Input Hysteresis EN Input Bias Current SW Switching Frequency SW Maximum Duty Cycle Error Amplifier Voltage Gain (4) Error Amplifier Transconductance COMP Maximum Output Current FB Regulation Threshold FB Input Bias Current SS Charging Current Symbol Condition VEN 0.3V VEN > 2V, VFB = 0.8V VIN = 2.6V to 25V VIN Rising 2.10 100 0.4 1.5 100 fSW VFB = 0.6V AVEA GEA Sourcing and Sinking 679 VFB = 0.7V During Soft-Start 0.9 85 1.1 93 400 350 30 700 2 1 1.3 Min Typ 0.5 0.9 2.4 Max 1.0 1.2 2.40 Units A mA V V mV V V mV A MHz % V/V A/V A mV A A
721 1
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER
ELECTRICAL CHARACTERISTICS (continued)
VIN = 5.0V, TA = +25C, unless otherwise noted.
Parameter SW On Resistance (4) SW Current Limit (4) SW Leakage Current Thermal Shutdown
(4)
Symbol Condition VIN = 5V VIN = 3V VSW = 25V Measured at OLS Pin
Min
3.0
Typ 150 225 4.0 0.5 160 27
Max
Units m m A A C V
Open Load Shutdown Threshold
Note: 4) Guaranteed by design.
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Load Current
100 FEEDBACK VOLTAGE (mV) 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 1 VIN = 12V 10 100 1000 LOAD CURRENT (mA) 10000
MP1517-EC01
Feedback Voltage vs Temperature
715 710 705 700 695 690 685 -50 50 100 0 TEMPERATURE (C) 150
VOUT = 18V
VOUT = 22V
MP1517-TPC01
Frequency vs Temperature
1.15 91.8 91.7 FREQUENCY (MHz) DUTY CYCLE (%) 1.10 91.6 91.5 91.4 91.3 91.2 1.00 -50 0 50 100 TEMPERATURE (C) 150
Maximum Duty Cycle vs Temperature
1.05
91.1 -50
0 50 100 TEMPERATURE (C)
150
MP1517-TPC02
MP1517-TPC03
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Operating Supply Current vs Temperature
940 920 900 880 860 840 820 -50 0 50 100 TEMPERATURE (C) 150 2.43 2.42
BP Voltage vs Temperature
4.8 4.6 4.4 4.2 4.0 3.8 3.6 3.4 3.2 0 50 100 TEMPERATURE (C) 150
Current Limit vs Temperature
BP VOLTAGE (V)
2.41 2.40 2.39 2.38 2.37 2.36 -50
CURRENT LIMIT (A)
3.0 -50
0 50 100 TEMPERATURE (C)
150
MP1517-TPC04
MP1517-TPC05
MP1517-TPC06
PIN FUNCTIONS
Pin # 1 2, 6, 14 3 4 5, 13 7 Name COMP NC BP EN SGND OLS Description Compensation: Error Amplifier Output. Connect to a series RC network to compensate the regulator control loop. No Connect Output of the Internal 2.4V Low Dropout Regulator. Connect a 10nF bypass capacitor between BP and SGND. Do not apply an external load to BP. Regulator On/Off Control Input. A logic high input (VEN > 1.5V) turns on the regulator, a logic low puts it into low current shutdown mode. The EN pin cannot be left floating. Signal Ground Open Load Shutdown Pin. OLS senses regulator output voltage to protect IC during open load operation. When this pin's voltage exceeds 27V, the output switch is shut off. The device then restarts in soft-start mode until it is disabled. Input Supply Pin. This pin can be connected to the regulator's input supply or to the output for boot-strapped operation. Output Switch Node. SW is the drain of the internal N-Channel MOSFET. Connect the inductor and rectifier to SW to complete the step-up converter. Power Ground Soft-Start Input. Connect a 10nF to 22nF capacitor from SS to SGND to set the soft-start time. SS sources 2A to an external soft-start capacitor during startup. As the voltage at SS increases to 0.55V, the voltage at COMP is clamped to 0.7V above the voltage at SS limiting the startup current. The external capacitor at SS is discharged to ground when under voltage lockout, thermal shutdown occurs or open load shutdown occurs. Regulation Feedback Input. The regulation threshold is 0.7V.
8 9, 10 11, 12
IN SW PGND
15
SS
16
FB
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER
OPERATION
The MP1517 uses a 1.1MHz fixed-frequency, current-mode regulation architecture to regulate the output. The operation of the MP1517 can be understood by referring to the block diagram of Figure 1. At the beginning of each cycle, the N-Channel MOSFET switch is turned on, forcing the inductor current to rise. The current at the source of the switch is internally measured and converted to a voltage by the current sense amplifier. That voltage is compared to the error voltage at COMP. The voltage at the output of the error amplifier is an amplified version of the difference between the
IN 8 BP EN OLS 3 4 7 16V 9, 10 SW PWM CONTROL LOGIC OSCILLATOR 2.4V LDO
700mV reference voltage and the feedback voltage. When these two voltages are equal, the PWM comparator turns off the switch forcing the inductor current to the output capacitor through the external rectifier. This causes the inductor current to decrease. The peak inductor current is controlled by the voltage at COMP, which in turn is controlled by the output voltage. Thus the output voltage controls the inductor current to satisfy the load. The use of current-mode regulation improves transient response and control loop stability.
+
0.9V
-+
2.4V -CURRENT SENSE AMP +
-SS 15 -SOFT-START GM + 0.7V
11, 12 PGND
16 FB
1
COMP
5, 13 SGND
MP1517_BD01
Figure 1--Functional Block Diagram
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER Internal Low-Dropout Regulator The internal power to the MP1517 is supplied from the IN pin through an internal 2.4V lowdropout linear regulator, whose output is BP. Bypass BP to SGND with a 10nF or greater capacitor to insure the MP1517 operates properly. The internal regulator can not supply any more current than is required to operate the MP1517, therefore do not apply any external load to BP. Soft-Start The MP1517 includes a soft-start timer that limits the voltage at COMP during startup to prevent excessive current at the input. This prevents premature termination of the source voltage at startup due to input current overshoot. When power is applied to the MP1517, and enable is asserted, a 2A internal current source charges the external capacitor at SS. As the capacitor charges, the voltage at SS rises. The MP1517 internally clamps the voltage at COMP to 700mV above the voltage at SS. This limits the inductor current at startup, forcing the input current to rise slowly to the current required to regulate the output voltage. Open Load Shutdown The MP1517 includes an open load detect that will stop the output from switching. In a fault condition where the connection to the LED's is open, VOUT will rise up. Once VOUT exceeds 27V, the MP1517 will stop switching and the output will stop rising. When the output falls below 27V the MP1517 will restart in soft-start mode and switches until the OLS threshold is exceeded again. This will continue until the part is disabled. To disable the open load shutdown feature, connect the OLS pin to GND.
APPLICATION INFORMATION
GENERAL PURPOSE COMPONENT SELECTION
Setting the Output Voltage Set the output voltage by selecting the resistive voltage divider ratio. Use 10k to 50k for the low-side resistor R2 of the voltage divider. Determine the high-side resistor R1 by the equation:
R1 = R2 x (VOUT - VFB ) VFB
conditions. A good rule for determining the inductance is to allow the peak-to-peak ripple current to be approximately 30% to 50% of the maximum input current. Make sure that the peak inductor current is below 3A to prevent loss of regulation due to the current limit. Calculate the required inductance value by the equation:
L= VIN x (VOUT - VIN ) VOUT x f SW x I
VOUT x ILOAD (MAX ) VIN x
where VOUT is the output voltage. For R2 = 10k and VFB = 0.7V, then R1 (k) = 14.29k (VOUT - 0.7V). Selecting the Inductor The inductor is required to force the higher output voltage while being driven by the input voltage. A larger value inductor results in less ripple current, resulting in lower peak inductor current and reducing stress on the internal N-Channel. MOSFET switch. However, the larger value inductor has a larger physical size, higher series resistance, and/or lower saturation current. Choose an inductor that does not saturate under the worst-case load transient and startup
MP1517 Rev. 1.4 4/28/2006
IIN(MAX ) =
I = (30% - 50%) x IIN(MAX )
Where VIN is the input voltage, fSW is the switching frequency, ILOAD(MAX) is the maximum load current, I is the peak-to-peak inductor ripple current and is the efficiency. Selecting the Input Capacitor An input capacitor is required to supply the AC ripple current to the inductor, while limiting noise at the input source. A low ESR capacitor is required to keep the noise at the IC to a minimum. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors may also suffice.
6
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER Use an input capacitor value greater than 10F. The capacitor can be electrolytic, tantalum or ceramic. However since it absorbs the input switching current it requires an adequate ripple current rating. Use a capacitor with RMS current rating greater than the inductor ripple current. To insure stable operation place the input capacitor as close to the IC as possible. Alternately a smaller high quality ceramic 0.1F capacitor may be placed closer to the IC with the larger capacitor placed further away. If using this technique, it is recommended that the larger capacitor be a tantalum or electrolytic type. All ceramic capacitors should be placed close to the MP1517. Selecting the Output Capacitor The output capacitor is required to maintain the DC output voltage. Low ESR capacitors are preferred to keep the output voltage ripple to a minimum. The characteristic of the output capacitor also affects the stability of the regulation control system. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended. In the case of ceramic capacitors, the impedance of the capacitor at the switching frequency is dominated by the capacitance, and so the output voltage ripple is mostly independent of the ESR. The output voltage ripple is calculated as:
VRIPPLE = ILOAD x (VO UT - VIN ) VO UT x C2 x f SW
design. Place the output capacitor close to SW to minimize the AC loop and switching noise. Selecting the Diode The output rectifier diode supplies current to the inductor when the internal MOSFET is off. To reduce losses due to diode forward voltage and recovery time, use a Schottky diode. Choose a diode whose maximum reverse voltage rating is greater than the maximum output voltage. The rated average forward current needs to be equal to or greater than the load current. Selecting the Soft-Start Capacitor The soft-start period is determined by the equation:
t SS = 0.275 x C4
Where CSS (in nF) is the soft-start capacitor from SS to SGND, and tSS (in ms) is the soft-start period. Determine the capacitor required for a given soft-start period by the equation:
C4 = 3.64 x t SS
It is recommended that values between 10nF and 22nF for CSS be used to set the soft-start period. Compensation The output of the transconductance error amplifier (COMP) is used to compensate the regulation control system. The system uses two poles and one zero to stabilize the control loop. The poles are fP1 set by the output capacitor and load resistance and fP2 set by the compensation capacitor C3. The zero fZ1 is set by the compensation capacitor C3 and the compensation resistor R3. These are determined by the equations: fP1 = fP2 =
1 x C2 x R LOAD G EA 2 x x C3 x A VEA 1 2 x x C3 x R3
Where VRIPPLE is the output ripple voltage, VIN and VOUT are the DC input and output voltages respectively, ILOAD is the load current, fSW is the switching frequency, and C2 is the capacitance of the output capacitor. In the case of tantalum or low-ESR electrolytic capacitors, the ESR dominates the impedance at the switching frequency, and so the output ripple is calculated as:
( VOUT - VIN ) x VOUT R + ESR VRIPPLE = ILOAD x VIN VOUT x f SW x C2
fZ1 =
Where RESR is the equivalent series resistance of the output capacitors. Choose an output capacitor to satisfy the output ripple and load transient requirements of the
Where RLOAD is the load resistance, GEA is the error amplifier transconductance, and AVEA is the error amplifier voltage gain.
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER The DC loop gain is: AVDC =
7 x A VEA x VIN x R LOAD x VFB V OUT
2
Where VFB is the feedback regulation threshold. There is also a right-half-plane zero (fRHPZ) that exists in the continuous conduction mode (inductor current does not drop to zero on each cycle) step-up converters. The frequency of the right-half plane zero is: fRHPZ =
VIN x R LOAD 2 x x L x VOUT
2 2
resistance (ESR) is used, then a second compensation capacitor (from COMP to SGND) is required to compensate for the zero introduced by the output capacitor ESR. The extra capacitor is required if the ESR zero is less than 4x the crossover frequency. The ESR zero frequency is: fZESR =
1 2 x x C2 x R ESR
If this is the case, calculate the second compensation capacitor by the equation: C6 =
C2 x R ESR R3
To stabilize the regulation control loop, the crossover frequency (The frequency where the loop gain drops to 0dB or a gain of 1) should be less than half of fRHPZ and should be at most 75KHz. fRHPZ is at its lowest frequency at maximum output load current. In some cases, an output capacitor with a high capacitance and high equivalent series
For most applications C6 is not required. Typical values for the compensation components are: C3 = 10nF R3 = 2.2k
SEPIC CONVERTER COMPONENT SELECTION
Selecting the Input Capacitor An input capacitor is required to supply the AC ripple current to the inductor, while limiting noise at the input source. The input capacitor selection is the same as that in the General Purpose Component Selection section above. Selecting the Inductors The SEPIC converter inductors (refer to Figure 4) are required to store energy, and generate an output voltage that is less than or greater than the input voltage. If a coupled inductor is used in a SEPIC converter, then the mutual inductance of each winding forces each inductor to become twice the original inductance. Therefore smaller inductance can be used with a coupled inductor. But the core saturation of the coupled inductors is related to the sum of both inductor currents. There are two current paths to the internal N-Channel MOSFET switch in a SEPIC converter. One is from L1 and the other is from L2. Each inductor's ripple current can be defined as:
L1 = L2 = D= VIN x D f SW x I
VOUT x (1 - D) f SW x I
VOUT + VD VOUT + VD + VIN
Where VD is the voltage drop on diode D1, and I is the peak to peak inductor ripple current. Set I to approximately 20% of the maximum switch current. Each inductor's peak current is:
IL1(PEAK ) = V + VD I + ILOAD x OUT 2 VIN V + VD I + ILOAD x IN 2 VIN
IL 2(PEAK ) =
The total of these two currents is the total switch current, and should be less than the minimum device current limit of 3A.
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MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER Selecting the Output Diode The output diode is typically a Schottky diode. The Schottky diode is selected based upon voltage requirement, current rating, and thermal capability. The diode breakdown and power switch breakdown voltage are set higher than VOUT + VIN, since this is the voltage stress on both of these devices. The current rating is set based on the average load current in the diode. The average diode current is equal to the load current, but the peak current and power consumption on D1 is:
ID1(PEAK ) = V + VD I + ILOAD x IN 2 VIN
Figure 3 shows a typical application running the MP1517 in flash mode. During preview mode, resistor R2 sets the current through the white LEDs to 20mA. When a flash is required the N-Channel MOSFET Q1 is turned on and 150mA flows through the LEDs. The compensation capacitor has been increased to 47nF. This forces the compensation node to slowly rise when Q1 is turned on which allows the current through the inductor to slowly increase without overshoot. By doing this the inrush current on the input is minimized. Figure 4 shows a SEPIC circuit using the MP1517 to generate 3.3V output from a 3V to 4.2V input. A peak voltage detect circuit (D2 and C7) is added to the IN pin from the highest voltage potential (SW node) in the entire circuit. The average voltage on C7 is roughly the sum of VIN and VOUT. This ensures the enhancement of the internal MOSFET switch. A transformer with two windings can be used to replace the two separate inductors in Figure 4. In this case, the effective inductance value is doubled due to the mutual inductance of each winding. This reduces the inductor ripple current and improves efficiency. Figures 5 and 6 show some other SEPIC application circuits. Layout Consideration High frequency switching regulators require very careful layout for stable operation and low noise. All components must be placed as close to the IC as possible. Keep the path between L1, D1, and C2 (also L2, C8 if applicable) extremely short for minimal noise and ringing. C1 must be placed close to the IN pin for best decoupling. All feedback components must be kept close to the FB pin to prevent noise injection on the FB pin trace. The ground return of C1 and C2 should be tied close to the PGND pins. See the MP1517 demo board layout for reference.
PD1 = ILOAD x VD
Selecting the Coupling Capacitor The steady state voltage across L1 and L2 is equal to zero. Therefore the coupling capacitor has VIN across it in steady state. The coupling capacitor will need to be rated for an input voltage plus some guard band. Also this capacitor will need an IRMS ripple current rating:
I C 8(RMS ) = ILOAD x VOUT + VD VIN
Application Examples Figure 2 shows a typical application circuit driving multiple strings of LEDs with the MP1517. The 3 strings of 6 white LEDs can be driven from a voltage supply range of 2.6V to 6V at an output current of 20mA. A 1F output capacitor is usually sufficient for this kind of application. A 4.7H inductor with low DCR (inductor resistance) is recommended to improve efficiency. A 10F ceramic capacitor is recommended for the input capacitance. Schottky diodes have fast recovery and a low forward voltage and are recommended. The MP1517 soft-start helps to limit the amount of current through VIN at startup and to also limit the amount of overshoot on the output.
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER
D1 MBR0540
VIN 2.6V to 20V
8 IN
OFF ON
4 15
EN SS
SW
9, 10 7 16 1
UP TO 6 LEDs PER STRING
OLS
C4 10nF C5 10nF
3
MP1517
BP FB
COMP SGND 5, 13 PGND 11, 12
C3 10nF
MP1517_F02
Figure 2--Driving 3 Strings of 6 White LEDs
VIN 2.6V to 6V
8
MBR0540 VOUT
SS OFF ON
4 15 3
IN SS EN SW 9, 10 7 16 1
OLS
MP1517
BP FB
COMP SGND 5, 13 PGND 11, 12
C3 47nF
Q1 NMOS ZXMN6A08E6TA
GATE
MP1517_F03
Figure 3--Flash Circuit Driving 4 White LEDs (150mA Flash Current)
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER
VIN 3V to 4.2V D2 1N4148 OR BAT54
8 IN
OFF ON
4 15
EN SS
SW
9, 10
D1 MBR320L
VOUT 3.3V/0.8A
C4 10nF C5 10nF
3 7
MP1517
BP OLS SGND 5, 13 FB
16 1
COMP PGND 11, 12
R3 OPEN C3 OPEN
C6 10nF
MP1517_F04
Figure 4--Typical SEPIC Application Circuit
VIN 3.3V D2 IN4148 OR BAT54
8 IN
OFF ON
4 15
EN SS
SW
9, 10
D1 MBR320L
VOUT 5V/0.6A
L2
C4 10nF C5 10nF
3 7
MP1517
BP OLS SGND 5, 13 FB
16 1
COMP PGND 11, 12
Q1 MMBT3904
C3 47nF
MP1517_F05
Figure 5--3.3V to 5V @ 600mA SEPIC Application Circuit
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER
VIN 5V
L1 VP1-0059 VERSA-PAC STANDARD MULTI-WINDING XFMR
8 IN
OFF ON
4 15
EN SS
SW
9, 10
D1 MBRS130
VOUT1 12V/0.2A
L2
C4 10nF C5 10nF
3 7
MP1517
BP OLS SGND 5, 13 FB
16 1
COMP PGND 11, 12
L3 C3 47nF D2 MBRS130
VOUT2 -12V/0.2A
MP1517_F06
Figure 6--12V SEPIC Application Circuit
MP1517 Rev. 1.4 4/28/2006
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MP1517 - 3A, 25V, 1.1MHz STEP-UP CONVERTER
PACKAGE INFORMATION
QFN16 (4mm x 4mm)
3.90 4.10 PIN 1 ID MARKING 0.25 0.35 PIN 1 ID INDEX AREA 3.90 4.10 0.65 BSC 9 4 0.50 0.70 13 2.15 2.45 16 PIN 1 ID SEE DETAIL A
12
1 2.15 2.45
8
5
TOP VIEW
BOTTOM VIEW
PIN 1 ID OPTION A 0.45x45 TYP. 0.80 1.00 0.00 0.05
PIN 1 ID OPTION B R0.25 TYP.
0.20 REF
SIDE VIEW
DETAIL A
3.80 2.30
NOTE:
1) ALL DIMENSIONS ARE IN MILLIMETERS. 2) EXPOSED PADDLE SIZE DOES NOT INCLUDE MOLD FLASH. 3) LEAD COPLANARITY SHALL BE 0.10 MILLIMETER MAX. 4) JEDEC REFERENCE IS MO-220, VARIATION VGGC. 5) DRAWING IS NOT TO SCALE.
1.00 0.35
0.65
RECOMMENDED LAND PATTERN
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications.
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